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19-0045; Rev 1; 5/94
In
it n K ded u atio alu n Incl Ev tio a form
Triple-Output Power-Supply Controller for Notebook Computers
____________________________Features
o o o o o o o Dual PWM Buck Controllers (+3.3V and +5V) Dual PCMCIA VPP Outputs (0V/3.3V/5V/12V) Two Precision Comparators or Level Translators Power-Ready Status Output (RDY5) 95% Efficiency Optimized for 6-Cell Applications 420A Quiescent Current; 70A in Standby (linear regulators alive) 25A Shutdown Current o 5.5V to 30V Input Range o Small SSOP Package o Fixed Output Voltages Available: 3.3V (standard) 3.45V (High-Speed PentiumTM) 3.6V (PowerPCTM)
_______________General Description
The MAX783 is a system-engineered power-supply controller for notebook computers or similar battery-powered equipment. It provides two high-performance step-down (buck) pulsewidth modulators (PWMs) for +3.3V/+5V and dual PCMCIA VPP outputs powered by an integral flyback winding controller. Other functions include dual, low-dropout, micropower linear regulators for CMOS/RTC back up, and two precision lowbattery-detection comparators. High efficiency (95% at 2A, greater than 80% at loads from 5mA to 3A) is achieved through synchronous rectification and PWM operation at heavy loads, and Idle-ModeTM operation at light loads. The MAX783 uses physically small components, thanks to high operating frequencies (300kHz/200kHz) and a new current-mode PWM architecture that allows for output filter capacitors as small as 30F per ampere of load. Line- and load-transient responses are terrific, with a high 60kHz unity-gain crossover frequency that allows output transients to be corrected within four or five clock cycles. Low system cost is achieved through a high level of integration and the use of low-cost external N-channel MOSFETs. The integral flyback winding controller provides a low-cost, +15V high-side output that regulates even in the absence of a load on the main output. Other features include low-noise, fixed-frequency PWM operation at moderate to heavy loads and a synchronizable oscillator for noise-sensitive applications such as electromagnetic pen-based systems and communicating computers. The MAX783 is similar to the MAX782, except the flyback winding is on the 3.3V inductor instead of the 5V inductor, the VPP outputs can be optionally programmed to 3.3V, and the device may be completely shut down.
MAX783
______________Ordering Information
PART MAX783CBX MAX783RCBX TEMP. RANGE 0C to +70C 0C to +70C PIN-PACKAGE 36 SSOP 36 SSOP VOUT 3.3V 3.45V
Ordering Information continued on last page.
__________________Pin Configuration
TOP VIEW
ON3 1 SHDN 2 D1 3 D2 4 VH 5 Q2 6 Q1 7 RDY5 8 VPPA 9 VDD 10 VPPB 11 36 SS3 35 CS3 34 FB3 33 DH3 32 LX3 31 BST3
________________________Applications
Notebook Computers Portable Data Terminals Communicating Computers Pen-Entry Systems
MAX783
30 DL3 29 V+ 28 VL 27 FB5 26 PGND 25 DL5 24 BST5 23 LX5 22 DH5 21 CS5 20 SS5 19 ON5
_______Typical Application Diagram
+3.3V 5.5V TO 30V VPP CONTROL ON3 ON5 SYNC SHDN 4 SUSPEND POWER POWER SECTION P MEMORY +5V PERIPHERALS
GND 12 REF 13 SYNC 14 DA1 15 DA0 16
MAX783
LOW-BATTERY WARNING VPP (0V/3.3V/5V/12V) VPP (0V/3.3V/5V/12V) DUAL PCMCIA SLOTS
DB1 17 DB0 18
SSOP
TMIdle-Mode is a trademark of Maxim Integrated Products. Pentium is a trademark of Intel. PowerPC is a trademark of IBM.
________________________________________________________________ Maxim Integrated Products 1
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
ABSOLUTE MAXIMUM RATINGS
V+ to GND .................................................................-0.3V, +36V PGND to GND........................................................................2V VL to GND ...................................................................-0.3V, +7V BST3, BST5 to GND ..................................................-0.3V, +36V LX3 to BST3.................................................................-7V, +0.3V LX5 to BST5.................................................................-7V, +0.3V Inputs/Outputs to GND ---- --- (D1, D2, SHDN, ON5, REF, SYNC, DA1, DA0, DB1, DB0, ON5, ---- --- SS5, CS5, FB5, RDY5, CS3, FB3, SS3, ON3).-0.3V, (VL + 0.3V) VDD to GND.................................................................-0.3V, 20V VPPA, VPPB to GND.....................................-0.3V, (VDD + 0.3V) VH to GND ...................................................................-0.3V, 20V Q1, Q2 to GND................................................-0.3V, (VH + 0.3V) DL3, DL5 to PGND..........................................-0.3V , (VL + 0.3V) DH3 to LX3 ..................................................-0.3V, (BST3 + 0.3V) DH5 to LX5 ..................................................-0.3V, (BST5 + 0.3V) REF, VL, VPP Short to GND........................................Momentary REF Current.........................................................................20mA VL Current ...........................................................................50mA VPPA, VPPB Current .........................................................100mA Continuous Power Dissipation (TA = +70C) SSOP (derate 11.76mW/C above +70C) ...................762mW Operating Temperature Ranges: MAX783CBX/MAX783_CBX.................................0C to +70C MAX783EBX/MAX783_EBX ..............................-40C to +85C Storage Temperature Range .............................-65C to +160C Lead Temperature (soldering, 10sec) .............................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
------- (V+ = 15V, GND = PGND = 0V, I VL = I REF = 0mA, S H D N = ON3 = ON5 = 5V, other digital input levels are 0V or +5V, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER 3.3V AND 5V STEP-DOWN CONTROLLERS Input Supply Range FB5 Output Voltage 0mV < (CS5-FB5) < 70mV, 6V < V + < 30V (includes load and line regulation) 0mV < (CS3-FB3) < 70mV, 6V < V + < 30V (includes load and line regulation) Either controller (0mV to 70mV) Either controller (6V to 30V) CS3-FB3 or CS5-FB5 CS3-FB3 (VDD < 13V, flyback mode) 80 -50 2.5 2 13 18 2 MAX783 MAX783R MAX783S CONDITIONS MIN 5.5 4.80 3.17 3.32 3.46 5.08 3.35 3.50 3.65 2.5 0.03 100 -100 4.0 TYP MAX 30 5.20 3.46 3.60 3.75 UNITS V V
ELECTRICAL CHARACTERISTICS
FB3 Output Voltage Load Regulation Line Regulation Current-Limit Voltage SS3/SS5 Source Current SS3/SS5 Fault Sink Current 15V FLYBACK CONTROLLER VDD Regulation Setpoint VDD Shunt Setpoint VDD Shunt Current Quiescent VDD Current Off VDD Current PCMCIA REGULATORS (Note 1)
V % %/V
120 -160 6.5
mV A mA V V mA A A
Falling edge, hysteresis = 1% Rising edge, hysteresis = 1% VDD = 20V VDD = 18V, ON3 = ON5 = 5V, VPPA/VPPB programmed to 12V with no external load VDD = 18V, ON3 = ON5 = 5V, VPPA/VPPB programmed to 0V Program to 12V, 13V < VDD < 19V, 0mA < IL < 60mA Program to 5V, 13V < VDD < 19V, 0mA < IL < 60mA Program to 3.3V, 13V < VDD < 19V, 0mA < IL < 60mA Program to 0V, 13V < VDD < 19V, 0mA < IL < 0.3mA
14 20 3 140 15 300 30
VPPA/VPPB Output Voltage
11.60 4.85 3.17 -0.30
12.10 5.05 3.30
12.50 5.20 3.43 0.30
V
2
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Triple-Output Power-Supply Controller for Notebook Computers
------- (V+ = 15V, GND = PGND = 0V, I VL = I REF = 0mA, S H D N = ON3 = ON5 = 5V, other digital input levels are 0V or +5V, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER CONDITIONS INTERNAL REGULATOR AND REFERENCE VL Output Voltage ON5 = ON3 = 0V, 5.5V < V+ < 30V, 0mA < IL < 25mA VL Fault Lockout Voltage Falling edge, hysteresis = 1% VL/FB5 Switchover Voltage Rising edge of FB5, hysteresis = 1% ---- --- (also RDY5 Trip Voltage) REF Output Voltage No external load (Note 2) REF Fault Lockout Voltage Falling edge REF Load Regulation 0mA < IL < 5mA (Note 3) --- - ---- SHDN = D1 = D2 = ON3 = ON5 = DA0 = DA1 = DB0 = Shutdown V+ Current DB1 = 0V, V+ = 30V D1 = D2 = ON3 = ON5 = DA0 = DA1 = DB0 = DB1 = 0V, Standby V+ Current V+ = 30V Quiescent Power Consumption D1 = D2 = D3 = DA0 = DA1 = DB0 = DB1 = 0V, (both controllers on) FB5 = CS5 = 5.25V, FB3 = CS3 = 3.5V Off V+ Current FB5 = CS5 = 5.25V, VL switched over to FB5 COMPARATORS D1, D2 Trip Voltage Falling edge, hysteresis = 1% D1, D2 Input Current D1 = D2 = 0V to 5V ---- --- Q1, Q2 RDY5 Source Current VH = 15V, VOUT = 2.5V ---- --- Q1, Q2 RDY5 Sink Current VH = 15V, VOUT 2.5V ---- --- Q1, Q2, RDY5 Output High Voltage ISOURCE = 5A, VH = 3V ---- --- Q1, Q2, RDY5 Output Low Voltage ISINK = 20A, VH = 3V Quiescent VH Current VH = 18V, D1 = D2 = 5V, no external load OSCILLATOR AND INPUTS/OUTPUTS SYNC = 3.3V Oscillator Frequency SYNC = 0V or 5V SYNC High Pulse Width SYNC Low Pulse Width SYNC Rise/Fall Time Not tested Oscillator SYNC Range SYNC = 3.3V Maximum Duty Cycle SYNC = 0V or 5V ---- --- Input Low Voltage SHDN, ON3, ON5, DA0, DA1, DB0, DB1, SYNC ---- --- SHDN, ON3, ON5, DA0, DA1, DB0, DB1 Input High Voltage SYNC ---- --- Input Current SHDN, ON3, ON5, DA0, DA1, DB0, DB1, VIN = 0V or 5V MIN 4.5 3.6 4.2 3.24 2.4 30 25 70 5.2 30 1.61 12 200 VH - 0.5 0.4 4 270 170 200 200 240 89 92 2.4 VL - 0.5 1 300 200 10 330 230 20 500 TYP MAX 5.5 4.2 4.7 3.36 3.2 75 40 110 8.6 60 1.69 100 30 1000 UNITS V V V V V mV A A mW A V nA A A V V A
ELECTRICAL CHARACTERISTICS (continued)
MAX783
kHz ns ns ns kHz %
200 350 92 95 0.8
V V A
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3
Triple-Output Power-Supply Controller for Notebook Computers MAX783
------- (V+ = 15V, GND = PGND = 0V, I VL = I REF = 0mA, S H D N = ON3 = ON5 = 5V, other digital input levels are 0V or +5V, TA = TMIN to TMAX, unless otherwise noted.) PARAMETER DL3/DL5 Sink/Source Current DH3/DH5 Sink/Source Current DL3/DL5 On-Resistance DH3/DH5 On-Resistance VOUT = 2V BST3-LX3 = BST5-LX5 = 4.5V, VOUT = 2V High or low High or low, BST3-LX3 = BST5-LX5 = 4.5V CONDITIONS MIN TYP 1 1 7 7 MAX UNITS A A
ELECTRICAL CHARACTERISTICS (continued)
Note 1: Output current is further limited by maximum allowable package power dissipation. Note 2: Because the reference uses VL as its supply, the REF line regulation error is insignificant. Note 3: The main switching outputs track the reference voltage. Loading the reference reduces the main outputs slightly according to the closed-loop gain (AVCL) and the reference voltage load regulation error. AVCL for the +3.3V supply is unity gain. AVCL for the +5V supply is 1.54.
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1, Transpower TTI5902 transformer, TA = +25C, unless otherwise noted.)
EFFICIENCY vs. +5V OUTPUT CURRENT
V+ = 6V V+ = 15V EFFICIENCY (%)
MAX183 5
EFFICIENCY vs. +3.3V OUTPUT CURRENT
MAX183 6
100
100 V+ = 6V 90 V+ = 15V 80
MAXIMUM +15V VDD OUTPUT CURRENT vs. SUPPLY VOLTAGE
500 MAXIMUM +15V LOAD CURRENT (mA) VDD > +13V +3.3V REGULATING 400 +3.3V LOAD = 0A 300 +3.3V LOAD = 3A
90 EFFICIENCY (%) 80
70 N1-N4 = IRF7101 ON3 = LOW F = 200kHz
70 N1-N4 = IRF7101 ON3 = ON5 = HIGH F = 200kHz
200
60
60
100
50 1m
10m
100m
1
10
50 1m
10m
100m
1
10
0 0 5 10 15 20 SUPPLY VOLTAGE (V)
+5V OUTPUT CURRENT (A)
+3.3V OUTPUT CURRENT (A)
QUIESCENT SUPPLY CURRENT vs. SUPPLY VOLTAGE
14 QUIESCENT SUPPLY CURRENT (mA) STANDBY SUPPLY CURRENT (mA) 2.5
STANDBY SUPPLY CURRENT vs. SUPPLY VOLTAGE
13 ON3 = ON5 = HIGH 2
2.0
1.5 ON3 = ON5 = 0V 1.0
1
0.5
0 0 6 12 18 24 30 SUPPLY VOLTAGE (V)
0 0 6 12 18 SUPPLY VOLTAGE (V) 24 30
4
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Triple-Output Power-Supply Controller for Notebook Computers
____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1, Transpower TTI5902 transformer, TA = +25C, unless otherwise noted.)
SHUTDOWN SUPPLY CURRENT vs. SUPPLY VOLTAGE
500 MINIMUM VIN TO VOUT DIFFERENTIAL (V) SHUTDOWN SUPPLY CURRENT (A) 1.0 SWITCHING FREQUENCY (kHz)
MAX783
MINIMUM VIN TO VOUT DIFFERENTIAL vs. +5V OUTPUT CURRENT
1000
SWITCHING FREQUENCY vs. LOAD CURRENT
SYNC = REF (300kHz) ON3 = ON5 = HIGH 100 +5V, VIN = 7.5V
400
0.8
300kHz
300 SHDN = 0V 200
0.6
10
+5V, VIN = 30V
0.4
200kHz +5V OUTPUT STILL REGULATING
1
+3.3V, VIN = 7.5V
100
0.2
0 0 6 12 18 24 SUPPLY VOLTAGE (V) 30
0 1m 10m 100m 1 10 +5V OUTPUT CURRENT (A)
0.1 100
1m
10m
100m
1
LOAD CURRENT (A)
PULSE-WIDTH MODULATION MODE WAVEFORMS
IDLE-MODE WAVEFORMS
LX VOLTAGE 10V/div
+5V OUTPUT 50mV/div
+5V OUTPUT VOLTAGE 50mV/div
2V/div
500ns/div ILOAD = 1A VIN = 16V ILOAD = 100mA VIN = 10V
200s/div
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5
Triple-Output Power-Supply Controller for Notebook Computers MAX783
____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1, Transpower TTI5902 transformer, TA = +25C, unless otherwise noted.) +5V LOAD-TRANSIENT RESPONSE +5V LINE-TRANSIENT RESPONSE, RISING
3A 0A
LOAD CURRENT
+5V OUTPUT 50mV/div
VIN, 10V TO 16V 2V/div +5V OUTPUT 50mV/div
200s/div VIN = 15V ILOAD = 2A
20s/div
+5V LINE-TRANSIENT RESPONSE, FALLING
+5V OUTPUT 50mV/div
VIN, 16V TO 10V 2V/div 20s/div ILOAD = 2A
OUTPUT NOISE SPECTRUM
OUTPUT VOLTAGE NOISE (dBmVRMS) 20 10 0 -10 -20 -30 VIN = 10V VOUT = 5V IOUT = 50mA OUTPUT VOLTAGE NOISE (dBmVRMS) 30 30 20 10 0 -10 -20 -30 -40 -50 -60 10k
OUTPUT NOISE SPECTRUM
VIN = 8V TO 12V (PLOTS SUPERIMPOSED) VOUT = 5V IOUT = 1A
1
10 FREQUENCY (kHz)
100
100k FREQUENCY (Hz)
1M
6
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Triple-Output Power-Supply Controller for Notebook Computers
______________________________________________________________Pin Description
PIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15-18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 NAME ON3 ---- --- SHDN D1 D2 VH Q2 Q1 ---- --- RDY5 VPPA VDD VPPB GND REF SYNC DA1, DA0, DB1, DB0 ON5 SS5 CS5 DH5 LX5 BST5 DL5 PGND FB5 VL V+ DL3 BST3 LX3 DH3 FB3 CS3 SS3 FUNCTION --- -- ON/OFF control input to disable the +3.3V PWM. Tie directly to VL for automatic start-up. Shutdown control input, low-true logic. Tie to VL for automatic start-up. The 5V VL supply stays active in ---- --- shutdown. Don't force SHDN higher than VL + 0.5V. #1 level-translator/comparator noninverting input, threshold = +1.650V. Controls Q1. Tie to GND if unused. #2 level-translator/comparator noninverting input (see D1). ---- --- External positive supply voltage input for the level translators/comparators and RDY5 output. #2 level-translator/comparator output. Sources 20A from VH when D2 is high. Sinks 500A to GND when D2 is low, even with VH = 0V. #1 level translator/comparator output (see Q2). Power-good indication for the main +5V supply. Low indicates greater than 4.5V at the +5V output. Swings 0V to VH. 0V, 3.3V, 5V, 12V switchable PCMCIA VPP output. Sources 60mA. Controlled by DA0 and DA1. +15V flyback input (feedback). A weak shunt regulator conducts 3mA to GND when VDD exceeds 19V. VDD serves as the supply input for the VPP linear regulators. 0V, 3.3V, 5V, 12V switchable PCMCIA VPP output. Sources 60mA. Controlled by DB0 and DB1. Low-current analog ground. Feedback reference point for all outputs. 3.3V reference output sources up to 5mA for external loads. Bypass to GND with 1F/mA of load or 0.22F minimum Oscillator control/synchronization input. Connect to VL or GND for 200kHz; connect to REF for 300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition starts a new cycle. PCMCIA digital control inputs with industry-standard coding (see Table 1). --- -- ON/OFF control input to disable the +5V PWM supply. Tie to VL for automatic start-up. Soft-start control input for +5V. Ramp time to full current limit is 1ms/nF of capacitance to GND. Current-sense input for +5V. Current limit level is +100mV referred to FB5. Gate-drive output for the +5V high-side MOSFET. Inductor connection for the +5V supply. Boost capacitor connection for the +5V supply (0.1F). Gate-drive output for the +5V low-side MOSFET. Power ground Feedback and current-sense input for the +5V PWM. 5V logic supply voltage for internal circuitry. VL is always on and can source 5mA for external loads. Supply voltage input from battery, 5.5V to 30V Gate-drive output for the +3.3V low-side MOSFET. Boost capacitor connection for the +3.3V supply (0.1F). Inductor connection for the +3.3V supply. Gate-drive output for the +3.3V high-side MOSFET. Feedback and current-sense input for the +3.3V PWM. Current-sense input for +3.3V, current limit level is +100mV referred to FB3. Soft-start input for +3.3V. Ramp time to full current limit is 1ms/nF of capacitance to GND.
MAX783
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7
Triple-Output Power-Supply Controller for Notebook Computers
Table 1. Truth Table for VPP Control Pins
D_0 0 0 1 1 D_1 0 1 0 1 VPP 0V 5V 12V 3.3V
_______________Detailed Description
The MAX783 converts a 5.5V to 30V input to six outputs (Figure 1). It produces two high-power, PWM switchmode supplies, one at +5V and the other at +3.3V. The two supplies operate at either 300kHz or 200kHz, allowing for small external components. Output current capability depends on external components, and can exceed 6A on each supply. Two 12V VPP outputs, an internal 5V, 25mA supply (VL) and a 3.3V, 5mA reference voltage are also generated via linear regulators (Figure 2). Faultprotection circuitry shuts off the PWM and high-side supply when the internal supplies lose regulation. Two precision voltage comparators are also included. Their output stages permit them to be used as level translators for driving external N-channel MOSFETs in load-switching applications, or for more conventional logic signals. The MAX783 is capable of accepting input voltages from 5.5V to 30V, but is optimized for the lower end of this range because the +15V flyback winding controller is appended to the +3.3V buck supply. This architecture allows for lower input voltages than are possible with the MAX782 sister chip, which puts the winding on the +5V side, while maintaining high +15V load capability. However, the MAX783's transformer has a higher turns ratio (4:1 vs. 2:1), which leads to higher interwinding capacitance as well as higher switching noise amplitudes at the transformer secondary when the input voltage is high. Therefore, the MAX783 standard application circuit is optimized with external components for low-voltage (6-8 cell) designs with maximum input voltages of 20V and less. The MAX783 itself can easily accept 30V inputs, but expect to see more noise and higher voltage swings at the transformer secondary under these conditions. The inductor and filter capacitor values may also require some adjustment for inputs greater than 20V; see the Design Procedure section.
cuit that uses a 100nF capacitor connected to BST5. The +5V supply's dropout voltage, as configured in Figure 1, is typically 400mV at 2A. As V+ approaches 5V, the +5V output falls with V+ until the VL regulator output hits its undervoltage lockout threshold at 4V. At this point, the +5V supply turns off. A synchronous rectifier at LX5 keeps efficiency high by effectively clamping the voltage across the rectifier diode. Maximum current limit is set by an external lowvalue sense resistor, which prevents excessive inductor current during start-up or under short-circuit conditions. Programmable soft-start is set by an optional external capacitor; this reduces in-rush surge currents upon start-up and provides adjustable power-up times for power-supply sequencing purposes.
MAX783
+3.3V Switch-Mode Supply
The +3.3V output is produced by a current-mode PWM step-down regulator similar to the +5V supply. The +3.3V supply uses a transformer primary winding as its inductor; the secondary is used for the 15V VDD supply. The default switching frequency for both PWM controllers is 200kHz (with SYNC connected to GND or VL), but 300kHz may be used by connecting SYNC to REF.
+3.3V and +5V PWM Buck Controllers
The two current-mode PWM buck controllers are nearly identical except for different preset output voltages and the addition of a flyback winding control loop to the 3.3V side. Each PWM is independent, except both are synchronized to a master oscillator and share a common reference (REF) and logic supply (VL). Each PWM can be turned on and off separately via ON3 and ON5. The PWMs are a direct-summing type, lacking a traditional integrator-type error amplifier and the phase shift associated with it. They therefore do not require external feedback compensation components if you follow the filter capacitor ESR guidelines in the Design Procedure. The main gain block is an open-loop comparator that sums four input signals: output voltage error signal, current-sense signal, slope-compensation ramp, and precision reference voltage. This direct-summing method approaches the ideal of cycle-by-cycle control of the output voltage. Under heavy loads, the controller operates in full PWM mode. Every pulse from the oscillator sets the output latch and turns on the high-side switch for a period determined by the duty factor (approximately VOUT/VIN). As the high-side switch turns off, the synchronous rectifier latch is set; 60ns later, the low-side switch turns on. The low-side switch stays on until the beginning of the next clock cycle (in continuous mode) or until the inductor current crosses through
+5V Switch-Mode Supply
The +5V supply is generated by a current-mode PWM step-down regulator using two N-channel MOSFETs, a rectifier, plus an LC output filter (Figure 1). The gatedrive signal to the high-side MOSFET, which must exceed the battery voltage, is provided by a boost cir8
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
BATTERY INPUT 5.5V TO 30V (NOTE 1)
C1 33F 16 VPP CONTROL INPUTS 15 18 17 DA0 DA1 DB0 DB1 VPPB VDD 29 V+ 28 VL VPPA 9 11 10 D1B BST3 DH3 LX3 DL3 CS3 31 33 32 30 35 N4 L2 10H D5 1N5819 R2 20m C7 4.7F C8 1F C9 1F C2 33F
+5V at 5mA
0V, 3.3V, 5V, 12V 0V, 3.3V, 5V, 12V +15V AT 200mA, SEE HIGH-SIDE SUPPLY (VDD) SECTION. C12 2.2F N3 1:4 D6 18V 100mW (NOTE 3) +3.3V at 3A
D1A C10 0.1F +5V at 3A R1 25m L1 10H D2 1N5819 N1 24 22 23 25 N2 21 27 (NOTE 2) C13 0.01F 20 1 19
MAX783
BST5 DH5 LX5 DL5 CS5 FB5 SS5 ON3 ON5
D3 EC11FS1
C11 0.1F
C3 C4 330F 330F +3.3V ON/OFF +5V ON/OFF
SHUTDOWN 5V POWERGOOD
2 SHDN 8 RDY5
FB3 34 36 SS3 5 VH 3, 4 D1-D2 7, 6 Q1-Q2 14 SYNC 13 REF
C14 0.01F 2 2
(NOTE 2)
C5 C6 150F 150F COMPARATOR SUPPLY INPUT COMPARATOR INPUTS COMPARATOR OUTPUTS OSCILLATOR SYNC 3.3V AT 5mA
PGND GND C1-C6 = SPRAGUE 595D or AVX TPS SERIES 26 12 N1-N4 = Si9410DY or IRF7101 (BOTH SECTIONS) D1A, D1B = LOW-POWER SCHOTTKY (CMPSH3 OR EQUIVALENT) FOR V+ < 6V. FOR V+ > 6V, 1N4148 OR EQUIVALENT IS ACCEPTABLE. NOTE 1: BATTERY VOLTAGE RANGE 6V to 20V WITH COMPONENTS SHOWN. NOTE 2: KEEP KELVIN-CONNECTED CURRENT-SENSE TRACES SHORT AND CLOSE TOGETHER. SEE FIG.5. NOTE 3: ZENER DIODE CLAMP REQUIRED FOR VIN > 12V. ZENER CAN BE REPLACED WITH 20k PULL-DOWN OR OTHER 1mA MINIMUM LOAD.
C15 1F
Figure 1. Standard Application Circuit
zero (in discontinuous mode). Under fault conditions when the inductor current exceeds the 100mV currentlimit threshold, the high-side latch resets and the highside switch turns off. At light loads, the inductor current fails to exceed the 25mV threshold set by the minimum current comparator. When this occurs, the PWM goes into idle mode, skipping most of the oscillator pulses in order to reduce the switching frequency and cut back switching losses. The oscillator is effectively gated off at light loads because the minimum current comparator immediately resets the high-side latch at the beginning of each cycle, unless the FB_ signal falls below the reference voltage level.
A flyback winding controller regulates the +15V VDD supply in the absence of a load on the main 3.3V output. If VDD falls below the preset +13V VDD regulation threshold, a 1s one-shot is triggered that extends the low-side switch's on-time beyond the point where the inductor current crosses zero (in discontinuous mode). This causes inductor (primary) current to reverse, pulling current out of the output filter capacitor and causing the flyback transformer to operate in the forward mode. The low impedance presented by the transformer secondary in forward mode allows the +15V filter capacitor to be quickly charged up again, bringing VDD into regulation.
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9
Triple-Output Power-Supply Controller for Notebook Computers MAX783
RDY5 V+ +5V LDO LINEAR REGULATOR 5V 4.5V ON
MAX783
P 5V PWM CONTROLLER (SEE FIG. 3) FB5 CS5 BST5 DH5 LX5 DL5 SS5
VL
3.3V REF GND
+3.3V REFERENCE
ON
PGND 4V SHDN FAULT ON5
2.8V STANDBY
SYNC
300kHz/200kHz ON OSCILLATOR
3.3V PWM CONTROLLER (SEE FIG. 3)
FB3 CS3 BST3 DH3 LX3 DL3 SS3
VPPA DA0 DA1
LINEAR REGULATOR
ON
VPPB DB0 DB1
LINEAR REGULATOR 13V VDD REG
ON3
VDD 13V TO 19V D1 Q1 1.65V D2 Q2 1.65V 19V
VH
Figure 2. Block Diagram
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
CS_ 1X
60kHz LPF REF, 3.3V (OR INTERNAL 5V REFERENCE) MAIN PWM COMPARATOR
FB_
R Q S LEVEL SHIFT
BST_ DH_ LX_
SLOPE COMP
OSC
MINIMUM CURRENT (IDLE-MODE) 25mV VL 4A CURRENT LIMIT SHOOTTHROUGH CONTROL
0mV-100mV SS_ 30R
3.3V ON_ N
1R
SYNCHRONOUS RECTIFIER CONTROL R -100mV N S VL Q LEVEL SHIFT DL_ PGND
N VDD REG (SEE FIG. 2) 1s SINGLE-SHOT
MAX783
Figure 3. PWM Controller Block Diagram
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Triple-Output Power-Supply Controller for Notebook Computers
Soft-Start/SS_ Inputs Connecting capacitors to SS3 and SS5 allows gradual build-up of the +3.3V and +5V supplies after ON3 and ON5 are driven high. When ON3 or ON5 is low, the appropriate SS capacitors are discharged to GND. When ON3 or ON5 is driven high, a 4A constant current source charges these capacitors up to 4V. The resulting ramp voltage on the SS_ pins linearly increases the current-limit comparator setpoint so as to increase the duty cycle to the external power MOSFETs up to the maximum output. With no SS capacitors, the circuit will reach maximum current limit within 10s. Soft-start greatly reduces initial in-rush current peaks and allows start-up time to be programmed externally. Synchronous Rectifiers Synchronous rectification allows for high efficiency by reducing the losses associated with the Schottky rectifiers. Also, the synchronous rectifier MOSFETS are necessary for correct operation of the MAX783's boost gate-drive and VDD supplies. When the external high-side power MOSFET turns off, energy stored in the inductor causes its terminal voltage to reverse instantly. Current flows in the loop formed by the inductor, Schottky diode, and load--an action that charges up the filter capacitor. The Schottky diode has a forward voltage of about 0.5V which, although small, represents a significant power loss and degrades efficiency. A synchronous rectifier MOSFET parallels the diode and is turned on by DL3 (or DL5) shortly after the diode conducts. Since the on resistance (rDS(ON)) of the synchronous rectifier is very low, the losses are reduced. The synchronous rectifier MOSFET is turned off when the inductor current falls to zero. Cross conduction (or "shoot-through") occurs if the highside switch turns on at the same time as the synchronous rectifier. Internal break-before-make timing ensures that shoot-through does not occur. The Schottky rectifier conducts during the time that neither MOSFET is on, which improves efficiency by preventing the synchronous-rectifier MOSFET's lossy body diode from conducting. The synchronous rectifier works under all operating conditions, including discontinuous-conduction and idle-mode. The +3.3V synchronous rectifier also controls the 15V VDD voltage (see the High-Side Supply (VDD) section). Boost Gate-Driver Supply Gate-drive voltage for the high-side N-channel switch is generated with a flying-capacitor boost circuit as shown in Figure 4. The capacitor is alternately charged from the VL supply via the diode and placed in parallel with
12
MAX783
BATTERY INPUT
VL VL BST_
LEVEL TRANSLATOR
DH_
PWM
VL
LX_
DL_
Figure 4. Boost Supply for Gate Drivers
the high-side MOSFET's gate-source terminals. On startup, the synchronous rectifier (low-side) MOSFET forces LX_ to 0V and charges the BST_ capacitor to 5V. On the second half-cycle, the PWM turns on the high-side MOSFET by connecting the capacitor to the MOSFET gate by closing an internal switch between BST_ and DH_. This provides the necessary enhancement voltage to turn on the high-side switch, an action that "boosts" the 5V gate-drive signal above the battery voltage. Ringing seen at the high-side MOSFET gates (DH3 and DH5) in discontinuous-conduction mode (light loads) is a natural operating condition caused by the residual energy in the tank circuit formed by the inductor and stray capacitance at the LX_ nodes. The gate driver negative rail is referred to LX_, so any ringing there is directly coupled to the gate-drive supply.
Modes of Operation
PWM Mode Under heavy loads--over approximately 25% of full load--the +3.3V and +5V supplies operate as continuous-current PWM supplies (see Typical Operating Characteristics). The duty cycle (%ON) is approximately: %ON = VOUT/VIN Current flows continuously in the inductor: First, it ramps up when the power MOSFET conducts; then, it ramps down during the flyback portion of each cycle as energy is put into the inductor and then discharged into the load. Note that the current flowing into the inductor when it is being charged is also flowing into the load, so the load is
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Triple-Output Power-Supply Controller for Notebook Computers
continuously receiving current from the inductor. This minimizes output ripple and maximizes inductor use, allowing very small physical and electrical sizes. Output ripple is primarily a function of the filter capacitor effective series resistance (ESR) and is typically under 50mV (see the Design Procedure section). Output ripple is worst at light load and maximum input voltage. Idle Mode Under light loads (<25% of full load), efficiency is further enhanced by turning the drive voltage on and off for only a single clock period, skipping most of the clock pulses entirely. Asynchronous switching, seen as "ghosting" on an oscilloscope, is thus a normal operating condition whenever the load current is less than approximately 25% of full load. At certain input voltage and load conditions, a transition region exists where the controller can pass back and forth from idle-mode to PWM mode. In this situation, short bursts of pulses occur that make the current waveform look erratic, but do not materially affect the output ripple. Efficiency remains high. ondary, as determined by the turns ratio. The secondary voltage is added to the +3.3V to make VDD. See the Typical Operating Characteristics for the VDD supply's load capability. Unlike other coupled-inductor flyback converters, the VDD voltage is regulated regardless of the loading on the +3.3V output. (Most coupled-inductor converters can only support the auxiliary output when the main output is loaded.) When the +3.3V supply is lightly loaded, the circuit achieves good control of VDD by pulsing the MOSFET normally used as the synchronous rectifier. This draws energy from the +3.3V supply's output capacitor and uses the transformer in a forwardconverter mode (i.e., the +15V output takes energy out of the secondary when current is flowing in the primary). These forward-converter pulses are interspersed with normal synchronous-rectifier pulses, and they only occur at light loads on the +3.3V rail. The transformer secondary's rectified and filtered output is only roughly regulated, and may be between 13V and 19V. It is brought back into VDD, which is also the feedback input, and used as the source for the PCMCIA VPP regulators. It can also be used as the VH power supply for the comparators or any external MOSFET drivers. When the input voltage is above 12V, or when the +3.3V supply is heavily loaded and VDD is lightly loaded, L2's interwinding capacitance and leakage inductance can produce voltages above that calculated from the turns ratio. A 2.5mA shunt regulator limits VDD to 19V. If the battery voltage can rise above 12V, VDD must either be externally clamped with an 18V zener diode, or there must be a 1mA minimum load on VDD (or VPPA/VPPB). Clock-frequency noise on the VDD rail of up to 3VP-P is a facet of normal operation, and can be reduced by adding more output capacitance.
MAX783
Current Limiting The voltage between CS3 (CS5) and FB3 (FB5) is continuously monitored. An external, low-value shunt resistor is connected between these pins, in series with the inductor, allowing the inductor current to be continuously measured throughout the switching cycle. Whenever this voltage exceeds 100mV, the drive voltage to the external high-side MOSFET is cut off. This protects the MOSFET, the load, and the battery in case of short circuits or temporary load surges. The current-limiting resistor R1 (R2) is typically 25m (20m) for a 3A load current. Oscillator Frequency; SYNC Input The SYNC input controls the oscillator frequency. Connecting SYNC to GND or to VL selects 200kHz operation; connecting to REF selects 300kHz operation. SYNC can also be driven with an external 240kHz to 350kHz CMOS/TTL source to synchronize the internal oscillator. 300kHz operation is used to minimize the inductor and filter capacitor sizes, but 200kHz may be necessary for low input voltages (see Low-Voltage Operation).
PCMCIA-Compatible, Programmable VPP Supplies
Two independent linear regulators furnish PCMCIA VPP supplies. The VPPA and VPPB outputs can be programmed to deliver 0V, 3.3V, 5V, or 12V. The 0V output mode has a 250 pull-down to discharge external filter capacitors and ensure that flash EPROMs cannot be accidentally programmed. These linear regulators draw their power from the high-side supply (VDD), and each can furnish up to 60mA. Bypass VPPA and VPPB to GND with at least 1F, with the bypass capacitors less than 20mm from the VPP pins. The outputs are programmed with DA0, DA1, DB0 and DB1, as shown in Table 2.
13
High-Side Supply (VDD)
The 15V VDD supply is obtained from the rectified and filtered secondary of transformer L2. VDD is enabled whenever the +3.3V supply is on (ON3 = high). The primary and secondary of L2 are connected so that, during the flyback (discharge) portion of each cycle, energy stored in the core is transferred into the +3.3V load through the primary and into VDD through the sec-
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Triple-Output Power-Supply Controller for Notebook Computers
Table 2. VPP Program Codes
DA0 0 0 1 1 DB0 0 0 1 1 DA1 0 1 0 1 DB1 0 1 0 1 VPPA 0V 5V 12V 3.3V VPPB 0V 5V 12V 3.3V
These codes are compatible with many popular PCMCIA digital controllers such as the Intel 82365SL. For other interfaces, one of the inputs can be permanently wired high or low and the other toggled to turn the supply on and off. The truth table shows that either a "0" or "1" can be used to turn each supply on. The two VPP outputs can be safely connected in parallel for increased load capability if the control inputs are also tied together (i.e., DA0 to DB0, DA1 to DB1). If VPAA and VPPB are connected in parallel, some devices may exhibit several milliamps of increased quiescent supply current when enabled, due to slightly mismatched output voltage set points.
The 3.3V precision reference (REF) is powered from the internal 5V VL supply. It can furnish up to 5mA for external loads. Bypass REF to GND with 0.22F, plus 1F/mA of load current. The main switch-mode outputs track the reference voltage. Loading the reference reduces the main output voltages slightly, according to the reference voltage load regulation error. Both the VL and REF supplies can remain active--even when the switch-mode regulators are turned off--to supply memory keep-alive power (see Shutdown Mode section). These linear regulator outputs can be directly connected to the corresponding switch-mode regulator outputs (i.e., REF to +3.3V, VL to +5V) to hold up the main supplies in standby mode. However, to ensure start-up, standby load currents must not exceed 5mA on each supply.
MAX783
Comparators
Two noninverting comparators can be used as precision voltage comparators or high-side drivers. The supply for these comparators (VH) is brought out and may be connected to any voltage between +3V and +19V. The noninverting inputs (D1-D2) are high impedance, and the inverting input is internally connected to a 1.650V reference. Each output (Q1-Q2) sources 20A from VH when its input is above 1.650V, and sinks 500A to GND when its input is below 1.650V. The Q1-Q2 outputs can be fixed together in wired-OR configuration since the pull-up current is only 20A. Connecting VH to a logic supply (5V or 3V) allows the comparators to be used as low-battery detectors. For driving N-channel power MOSFETs to turn external loads on and off, VH should be 6V to 12V higher than the load voltage. This enables the MOSFETs to be fully turned on and results in low rDS(ON). VDD is a convenient source for VH.
Shutdown Mode ---- --- Shutdown (SHDN = low) forces both PWMs off and disables the -- -- output and the auxiliary comparators REF- -- including RDY5. Supply current in shutdown mode is typically 25A. The VL supply remains active and can source 25mA for external loads. VL load capability is higher in shutdown and standby modes than when the PWMs are operating (25mA vs. 5mA). Standby mode -is achieved by holding ON3 and ON5 ---- -- low while SHDN is high. This disables both PWMs, but keeps VL, REF, and the precision comparators alive. Supply current in standby mode is typically 70A. Other ways to shut down the MAX783 are suggested in the applications section of the MAX782 data sheet.
__________________Design Procedure
Figure 1's predesigned application circuit contains the correct component values for 3A output currents and a 6V to 20V input range. Use the design procedure that follows to optimize this basic schematic for different voltage or current requirements. Before beginning a design, firmly establish the following: VIN(MAX), the maximum input (battery) voltage. This value should include the worst-case conditions under which the power supply is expected to function, such as no-load (standby) operation when a battery charger is connected but no battery is installed. VIN(MAX) cannot exceed 30V. VIN(MIN), the minimum input (battery) voltage. This value should be taken at the full-load operating current under the lowest battery conditions. If VIN(MIN) is below about 6V, the filter capacitance required to maintain good AC load regulation increases, and the current limit for the +5V supply has to be increased for the same load level.
Internal VREF and VL Supplies
An internal linear regulator produces the 5V used by the internal control circuits. This regulator's output is available on pin VL and can source 5mA for external loads. Bypass VL to GND with 4.7F. To save power, when the +5V switch-mode supply is above 4.5V, the VL linear regulator is turned off and the high-efficiency +5V switch-mode supply output is internally connected to VL.
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Triple-Output Power-Supply Controller for Notebook Computers
+5V Inductor (L1)
Three inductor parameters are required: the inductance value (L), the peak inductor current (ILPEAK), and the coil resistance (RL). The inductance is: (VOUT) (VIN(MAX) - VOUT) L = ---------------------- (VIN(MAX)) (f) (IOUT) (LIR) VOUT = output voltage, 5V VIN(MAX) = maximum input voltage (V) f = switching frequency, normally 300kHz IOUT = maximum +5V DC load current (A) LIR = ratio of inductor peak-to-peak AC current to average DC load current, typically 0.3. A higher value of LIR allows smaller inductance, but results in higher losses and higher ripple. The highest peak inductor current (ILPEAK) equals the DC load current (IOUT) plus half the peak-to-peak AC inductor current (ILPP). The peak-to-peak AC inductor current is typically chosen as 30% of the maximum DC load current, so the peak inductor current is 1.15 times IOUT. The peak inductor current at full load is given by: (VOUT) (VIN(MAX) - VOUT) ILPEAK = IOUT + ---------------------- (2) (f) (L) (VIN(MAX)) The coil resistance should be as low as possible, preferably in the low milliohms. The coil is effectively in series with the load at all times, so the wire losses alone are approximately: Power loss = (IOUT2)(RL) In general, select a standard inductor that meets the L, ILPEAK, and RL requirements (see Tables 3 and 4). If a standard inductor is unavailable, choose a core with an LI2 parameter greater than (L)(ILPEAK2), and use the largest wire that will fit the core. where: primary. VDD current (IDD) usually includes the VPPA and VPPB output currents. The total +3.3V power, PTOTAL, is the sum of these powers: PTOTAL = P3 + PDD where: P3 = (VOUT) (IOUT); PDD = (VDD) (IDD); and: VOUT = output voltage, 3.3V; IOUT = maximum +3.3V load current (A); VDD = VDD output voltage, 15V; IDD = maximum VDD load current (A); so: PTOTAL = (3.3V x IOUT) + (15V x IDD) and the equivalent +3.3V output current, ITOTAL, is: ITOTAL = PTOTAL / 3.3V = [(3.3V x IOUT) + (15V x IDD)] / 3.3V. The primary inductance, LP, is given by: (VOUT) (VIN(MAX) - VOUT) LP = -------------------------- (VIN(MAX)) (f) (ITOTAL) (LIR) where: VIN(MAX) = maximum input voltage f = switching frequency, normally 300kHz ITOTAL = maximum equivalent load current (A) LIR = ratio of primary peak-to-peak AC current to average DC load current, typically 0.3. The highest peak primary current (I LPEAK) equals the total DC load current (ITOTAL) plus half the peak-to-peak AC primary current (ILPP). The peak-to-peak AC primary current is typically chosen as 30% of the maximum DC load current, so the peak primary current is 1.15 times ITOTAL. A higher value of LIR allows smaller inductance, but results in higher losses and higher ripple. The peak current in the primary at full load is given by: (VOUT) (VIN(MAX) - VOUT) ILPEAK = ITOTAL + --------------------------. (2) (f) (LP) (VIN(MAX)) Choose a core with an LI 2 parameter greater than (LP) (ILPEAK2). The winding resistances, RP and RS, should be as low as possible, preferably in the low milliohms. Use the largest gauge wire that will fit on the core. The coil is effectively in series with the load at all times, so the resistive losses in the primary winding alone are approximately (ITOTAL)2 (RP). The minimum turns ratio, NMIN, is 3.3V:(15V-3.3V). Use 1:4 to accommodate the tolerance of the +3.3V supply. A greater ratio will reduce efficiency of the VPP regulators. Minimize the diode capacitance and the interwinding capacitance, since they create losses through the VDD shunt regulator. These are most significant when the input voltage is high, the +3.3V load is heavy, and there is no load on VDD.
15
MAX783
+3.3V Transformer (L2)
Table 3 lists two commercially available transformers and parts for a custom transformer. The following instructions show how to determine the transformer parameters required for a custom design: LP, the primary inductance value ILPEAK, the peak primary current LI2, the core's energy rating RP and RS, the primary and secondary resistances N, the primary-to-secondary turns ratio. The transformer primary is specified just as the +5V inductor, using VOUT = +3.3V; but the secondary output (VDD) power must be added in as if it were part of the
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
FAT, HIGH-CURRENT TRACES MAIN CURRENT PATH
about twice the value of the sense resistor. MOSFETs with even lower rDS(ON) have higher gate capacitance, which increases switching time and transition losses. MOSFETs with low gate-threshold voltage specifications (i.e., maximum VGS(TH) = 2V rather than 3V) are preferred, especially for high-current (5A) applications.
KELVIN SENSE TRACES
SENSE RESISTOR
Output Filter Capacitors (C3-C6)
The output filter capacitors determine the loop stability and output ripple voltage. To ensure stability, the minimum capacitance and maximum ESR values are: VREF CF > -------------------------- (VOUT) (RCS) (2) () (GBWP) and, (VOUT) (RCS) ESRCF < ------------ VREF CF = output filter capacitance, C6 or C7 (F) VREF = reference voltage, 3.3V VOUT = output voltage, 3.3V or 5V RCS = sense resistor () GBWP = gain-bandwidth product, 60kHz ESRCF = output filter capacitor ESR (). Be sure to select output capacitors that satisfy both the minimum capacitance and maximum ESR requirements. To achieve the low ESR required, it may be appropriate to use a capacitance value 2 or 3 times larger than the calculated minimum. The output ripple in continuous-current mode is: VOUT(RPL) = (ILPP(MAX)) [(ESRCF+1/(2 x x f x CF)]. In idle-mode, the ripple has a capacitive and resistive component: (4 x 10-4) (L) VOUT(RPL)(C) = -------------- x (RCS2) (CF) 1 1 ------ + ---------- Volts VOUT VIN - VOUT where:
MAX783
Figure 5. Kelvin Connections for the Current-Sense Resistors
Ensure the transformer secondary is connected with the right polarity: A VDD supply will be generated with either polarity, but proper operation is possible only with the correct polarity. Test for correct connection by observing the phase relationship between the LX3 switching node and the transformer secondary under load. The two waveforms must be 180 out of phase.
Current-Sense Resistors (R1, R2)
The sense resistors must carry the peak current in the inductor, which exceeds the full DC load current. The internal current limiting starts when the voltage across the sense resistors exceeds 100mV nominally, 80mV minimum. Use the minimum value to ensure adequate output current capability: For the +5V supply, R1 = 80mV / (1.15 x I OUT); for the +3.3V supply, R2 = 80mV/(1.15 x ITOTAL), assuming that LIR = 0.3. Since the sense resistance values (e.g., R1 = 25m for IOUT = 3A) are similar to a few centimeters of narrow traces on a printed circuit board, trace resistance can contribute significant errors. To prevent this, Kelvin connect the CS_ and FB_ pins to the sense resistors; use separate traces not carrying any of the inductor or load current, as shown in Figure 5. Run these traces parallel at minimum spacing from one another. The wiring layout for these traces is critical for stable, lowripple outputs (see the Layout and Grounding section).
(
)
MOSFET Switches (N1-N4)
The four N-channel power MOSFETs are usually identical and must be "logic-level" FETs; that is, they must be fully on (have low rDS(ON)) with only 4V gate-source drive voltage. The MOSFET rDS(ON) should ideally be
(0.02) (ESRCF) VOUT(RPL)(R) = ------------- Volts RCS The total ripple, VOUT(RPL), can be approximated as follows: if VOUT(RPL)(R) < 0.5 VOUT(RPL)(C), then VOUT(RPL) = VOUT(RPL)(C), otherwise, VOUT(RPL) = 0.5 VOUT(RPL)(C) + VOUT(RPL)(R).
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Triple-Output Power-Supply Controller for Notebook Computers
Diode D3
The voltage rating of D3 should be at least 4 x VIN + 5V plus a safety margin. A rating of at least 100V is necessary for the maximum 20V supply. Use a highspeed silicon diode (with a higher breakdown voltage and low capacitance) rather than a Schottky diode. D3's current rating should exceed twice the maximum current load on VDD. of the MAX783 circuit does not exceed the bypass capacitor ripple current rating. The RMS input current (IRMS) can be calculated as shown below: IRMS = RMS AC input current ---------------- VOUT (VIN - VOUT) = ILOAD ------------------- VIN
MAX783
(
)
Diodes D2 and D5
Use 1N5819s or similar Schottky diodes. D2 and D5 conduct only about 3% of the time, so the 1N5819's 1A current rating is conservative. The voltage rating of D2 and D5 must exceed the maximum input supply voltage from the battery. These diodes must be Schottky diodes to prevent the lossy MOSFET body diodes from turning on, and they must be placed physically close to their associated synchronous rectifier MOSFETs.
Low-Voltage Operation
Low input voltages, such as the 6V end-of-life voltage of a 6-cell NiCd battery, place extra demands on the +5V buck regulator because of the very low input-output differential voltage. The standard application circuit works well with supply voltages down to 6V; at input voltages less than 6V, the +5V filter capacitor values must be increased. If the minimum battery voltage is 6.5V or higher, the 660F total 5V filter capacitance can be reduced to 330F. The +5V supply's load-transient response is impaired due to reduced inductor-current slew rate, which is in turn caused by reduced voltage applied across the buck inductor during the high-side switch-on time. So, the +5V output sags when hit with an abrupt load current change, unless the +5V filter capacitor value is increased. Only the capacit a n c e i s a f f e c t e d a n d E S R r e q u i r e m e n t s d o n 't change. Therefore, the added capacitance can be supplied by an additional low-cost bulk capacitor in parallel with the normal low-ESR switching-regulator capacitor. The equation for voltage sag under a step-load change follows: (ISTEP2)(L) VSAG = ---------------------------------- (2)(CF) (VIN(MIN) x DMAX - VOUT) where DMAX is the maximum duty cycle. Higher duty cycles are possible when the oscillator frequency is reduced to 200kHz, due to fixed propagation delays through the PWM comparator becoming a lesser part of the whole period. The tested worst-case limit for DMAX is 92% at 200kHz. Lower inductance values can reduce the filter capacitance requirement, but only at the expense of increased noise at high input voltages (resulting from higher peak currents).
Soft-Start Capacitors (C13, C14)
A capacitor connected from GND to either SS pin causes that supply to ramp up slowly. The ramp time to full current limit, tSS, is approximately 1ms for every nF of capacitance on SS_, with a minimum value of 10s. Typical capacitor values are in the 10nF to 100nF range. Because this ramp is applied to the current-limit circuit, the actual time for the output voltage to ramp up depends on the load current and output capacitor value. Using Figure 1's circuit with a 2A load and no SS capacitor, full output voltage is reached in less than 1ms after ON_ is driven high.
Bypass Capacitors
Input Filter Capacitors (C1, C2) Use at least 3F/W of output power for the input filter capacitors, C1 and C2. They should have less than 150m ESR, and should be located no further than 10mm from N1 and N2 to prevent ringing. Connect the negative terminals directly to PGND. Be careful not to exceed the surge current ratings of the bypass capacitors. If the battery pack or AC adapter has very low output impedance, tantalum capacitors may be damaged when initial connection is made. In this situation, electrolytic capacitors such as Sanyo OS-CON may be necessary. Also, take care that the RMS input current
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17
Triple-Output Power-Supply Controller for Notebook Computers MAX783
Layout and Grounding
Good layout is necessary to achieve the designed output power, high efficiency, and low noise. Good layout includes use of a ground plane, appropriate component placement, and correct routing of traces using appropriate trace widths. The following points are in order of importance: 1. A ground plane is essential for optimum performance. In most applications, the power supply is located on a multilayer motherboard, and full use of the four or more copper layers is recommended. Use the top and bottom layers for interconnections, and the inner layers for an uninterrupted ground plane. 2. Keep the Kelvin-connected current-sense traces short, close together, and away from switching nodes. See Figure 5. Important: Place the currentsense resistors close to the IC (less than 10mm away if possible). 3. Place the LX node components N1, N2, D2, and L1 as close together as possible. This reduces resistive and switching losses and keeps noise due to ground inductance confined. Do the same with the other LX node components N3, N4, D5, and L2. 4. The input filter capacitor C1 should be less than 10mm away from N1's drain. The connecting copper trace carries large currents and must be at least 2mm wide, preferably 5mm. Similarly, place C2 close to N3's drain, and connect them with a wide trace. 5. Keep the gate connections to the MOSFETs short for low inductance (less than 20mm long and more than 0.5mm wide) to ensure clean switching. 6. To achieve good shielding, it is best to keep all high-voltage switching signals (MOSFET gate drives DH3 and DH5, BST3 and BST5, and the two LX nodes) on one side of the board and all sensitive nodes (CS3, CS5, FB3, FB5 and REF) on the other side. 7. Connect the GND and PGND pins directly to the ground plane, which should ideally be an inner layer of a multilayer board. 8. Connect the bypass capacitor C7 very close (less than 10mm) to the VL pin. 9. Minimize the capacitance at the transformer secondary. Place D3 and C12 very close to each other and to the secondary, then route the output to the IC's VDD pin with a short trace. Bypass with 0.1F close to the VDD pin if this trace is longer than 50mm. The layout for the evaluation board is shown in the Evaluation Kit section. It provides an effective, lownoise, high-efficiency example.
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
Table 3. Surface-Mount Components
(See Figure 1 for schematic and Table 4 for manufacturers' telephone numbers.) COMPONENT C1, C2 C3, C4 C5, C6 C7 C8, C9, C15 C10, C11 C12 C13, C14 D1 D3 D2, D5 D6 R1 R2 N1-N4 L1 L2 TYPE 33F, 35V tantalum 33F, 25V tantalum 330F, 10V tantalum 330F, 6.3V tantalum 150F, 10V tantalum 220F, 10V tantalum 4.7F 16V tantalum 1F, 20V tantalum 0.1F, 16V tantalum 2.2F, 25V tantalum 0.01F, ceramic Dual low-power Schottky Fast silicon rectifier 1N5819 Schottky 18V, 100mW zener diode 0.025 SMT resistor 0.02 SMT resistor N-channel MOSFETs 10H, 2.5A inductor 10H, 1:4 transformer (Note 2) MANUFACTURER Sprague AVX Sprague AVX Sprague AVX Sprague Sprague Murata-Erie Sprague Murata-Erie Central Semiconductor Nihon Nihon Central Semiconductor IRC IRC International Rectifier Sumida Transpower Transpower Coiltronics PART NUMBER 595D336X0035R2B TPSE336M025R0300 595DD337X0010R2B TPSE337M006R0100 595D157X0010D2B TPSE227M010R0100 595D475X0016A 595D105X0020T GRM42-6X7R104K50V 595D225X0025B GRM42-6X7R103K50V CMPSH-3A EC11FS1 EC10QS04 CMPZ5248 LR2010-01-R025-F LR2010-01-R020-F IRF7101 (Note 1) CDR125-100 TTI5897 (for 3.3V at 1A) TTI5902 (for 3.3V at 3A) CTX03-12210 (for 3.3V at 2A)
Note 1: Four IRF7101s total; each device has both sections connected in parallel. Note 2: These transformers have different sizes and pinouts. The MAX783 EV kit has the correct pad layout for the TTI5902 transformer, but all the transformers listed can be wired in easily.
Table 4. Surface-Mount Component Suppliers
Company AVX Central Semi Coiltronics International Rectifier IRC Murata-Erie Nihon Sprague Sumida Transpower Tech. *Distributor ______________________________________________________________________________________ 19 Factory Fax [country code] [ 1 ] 207-283-1941 [ 1 ] 516-435-1824 [ 1 ] 407-241-9339 [ 1 ] 310-322-3332 [ 1 ] 512-992-3377 [ 1 ] 404-736-3030 [81] 3-3494-7414 [ 1 ] 508-339-5063 [81] 3-3607-5428 [ 1 ] 702-831-3521 USA Phone (207) 282-5111 (800) 282-4975 (516) 435-1110 (407) 241-7876 (310) 322-3331 (512) 992-7900 (404) 736-1300 (805) 867-2555* (508) 339-8900 (708) 956-0666 (702) 831-0140
Triple-Output Power-Supply Controller for Notebook Computers MAX783
______________________________________________Evaluation Kit Information ___________EV Kit Standard Features
o Battery Range: 5.5V to 20V* o Load Capability: 5V at 3A* 3.3V at 3A* 12V at 120mA or 15V at 200mA o 3.3V and 5V Keep-Alive Linear Regulator Outputs o Dual PCMCIA VPP Outputs o Oscillator Sync Input *For wider input voltage range or higher load current, see the Design Procedure. R12 R13 R14 Open-circuit (no resistor installed) 100k 5% chip resistor (SYNC pull-down, usually shorted out) 560 5% resistor (pull-up for comparator supply, usually shorted out or left open)
___________EV Kit Quick Reference
Connect a stiff (30W or better) bench power supply to the +VIN and GND pads found on the edge of the board. Turn up the input voltage to somewhere between 5V and 20V. Set switches SW2A, SW2B, and SW2C on (if they are not already on), taking the device out of shutdown and turning on the main switching regulators. Set switch SW2D off so that the oscillator is set to 200kHz, which is the appropriate frequency for the 6V-20V input range. The main outputs are now regulating and ready for heavy loads. For best output accuracy and noise characteristics, loads should be returned separately to the GND pad corresponding to and adjacent to each output (+3OUT and +5OUT). Normal full-load regulation error is typically -2.5% while keeping the outputs within tolerance. If the measured error is higher, there may be drops in the wiring or ground. Ensure the voltmeter is sensing directly at the output and ground pads. To observe normal PWM switching action, place a 1A load on either output and observe the corresponding switching node (device LX_ pin) while varying the input voltage. Without a load, the switching waveforms are intermittent and difficult to trigger on and it may appear that the board isn't working (except for the presence of output voltage). To exercise the VPP controls, first ensure the 3.3V switch-mode supply is ON. The loosely regulated flyback voltage (13V to 19V) will be present at the edge pad marked "+15OUT". Turn on each VPP output by setting the appropriate code on SW1, where ON = HIGH (see Table 1). The 5V linear regulator output is always present, even in shutdown mode, and can be measured at the VL pad. If the battery is connected and VL is not at 5V or close to it, something is wrong (excess VL load, possibly). The precision 3.3V linear regulator (REF) is activated by taking the device out of shutdown mode (by turning SW2C on). Turning on the SW2D sets the oscillator to 300kHz, but the external component values should be changed first (see the Design Procedure). The oscillator can be synchronized to an external clock signal by driving the SYNC pad with a 240kHz to 350kHz pulse train of 5V amplitude while SW2D is off.
_________________EV Kit Description
The MAX783 evaluation kit (EV kit) is a preassembled and tested demonstration board that embodies the standard application circuit, with some extra pull-up and pull-down resistors needed to set default logic signal levels. The board comes configured to accept battery voltages between 5.5V and 20V, but it can be reconfigured for voltages between 5.5V and 30V. The maximum voltage for safe operation (20V) is determined by the breakdown voltage rating of the external MOSFETs and input filter capacitor (C1 and C2) voltage ratings; if these are replaced with high-voltage devices, the board can tolerate 30V input without damage (36V absolute max). Load current capability of the standard board is 3A at each main output (5V and 3.3V) and 60mA at each 12V VPP output. Load current capability can also be configured by selecting appropriate external components and sense resistor values, with practical load capability up to 7A at each main output and 500mA on the 15V flyback output (see the Design Procedure for reconfiguring both load and input voltage capability). All functions are controlled by two on-board dipswitches, which can be overridden by external CMOS/TTL logic signals if desired (provided the dipswitches are set to off first). See Table 3 for the EV kit components. These components correspond exactly to Figure 1's Standard Application Circuit, with the addition of the following chip resistors (which are required only to set default logic and supply levels). The comparator outputs both swing 0V to 5V unless R14 is removed and R12 is installed, in which case their output swing is 0V to 15V. R3-R11 1M 5% resistors (logic pull-down, usually not needed and not installed in normal applications)
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+VIN C2 33F VL 29 V+ DA1 VPPA C8 1F VPPB 11 VPPB C9 1F VDD 10 D1B 24 BST5 BST3 D3 +15OUT 33 N3 R2 20m L2 10H N4 C5 150F +3.3OUT C6 150F C11 32 0.1F C12 2.2F D6 18V 31 DA0 DB1 DB0 C7 4.7F A SW1 15 16 17 18 R3 D1A R4 R5 R6 28 VL VPPA 9 B C C1 33F D
DA1
DA0
Figure 6. MAX783 EV Kit Schematic
R1 25m C4 330F LX3 30 D5 D2 21 CS5 CS3 34 TO VL SS3 C14 36 0.01F R12 OPEN VH RDY5 Q1 Q2 4 R7 R8 R9 R10 R11 PGND 26 D2 SYNC 5 8 7 6 3 D1 14 13 REF GND 12 C15 1F R13 100k SW2D R14 560 VH RDY5 Q1 Q2 SYNC REF 35 N2 25 DL5 DL3 L1 10H C10 0.1F 23 LX5 N1 DH5 DH3 22
DB1
DB0
MAX783
+5OUT
C3 330F
27 FB5 FB3 A SW2 1 ON3 19 ON5 2 SHDN B C13 C 0.01F 20 SS5
ON3
ON5
SHDN
D1
MAX783
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R3 - R11 = 1M
D2
Triple-Output Power-Supply Controller for Notebook Computers
21
Triple-Output Power-Supply Controller for Notebook Computers MAX783
Figure 7. MAX783 EV Kit Top Component Layout and Silk Screen, Top View
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
Figure 8. MAX783 EV Kit Ground Plane (Layers 2 and 3), Top View
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23
Triple-Output Power-Supply Controller for Notebook Computers MAX783
Figure 9. MAX783 EV Kit Top Layer (Layer 1), Top View
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
Figure 10. MAX783 EV Kit, Bottom Component Layout and Silk Screen, Bottom View
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25
Triple-Output Power-Supply Controller for Notebook Computers MAX783
Figure 11. MAX783 EV Kit, Bottom Layer (Layer 4), Top View
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
Figure 12. MAX783 EV Kit, Drill Schedule
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Triple-Output Power-Supply Controller for Notebook Computers MAX783
___________________Chip Topography
D2 SHDN SS3 FB3 DH3 CS3 D1 ON3
VH Q2 Q1 RDY5 VPPA LX3 BST3
_Ordering Information (continued)
PART MAX783SCBX MAX783C/D MAX783EBX MAX783REBX MAX783SEBX EV KIT MAX783EVKIT-SO TEMP. RANGE 0C to +70C 0C to +70C -40C to +85C -40C to +85C -40C to +85C TEMP. RANGE 0C to +70C PIN-PACKAGE 36 SSOP Dice* 36 SSOP 36 SSOP 36 SSOP VOUT 3.6V -- 3.3V 3.45V 3.6V
DL3 VDD 0.181" VL (4.597mm) FB5 PGND DL5 V+
BOARD TYPE Surface Mount
VPPB GND REF SYNC DA0 DB0 DA1 DB1 SS5 ON5 CS5 DH5
* Dice are specified at +25C only.
BST5 LX5
0.132" (3.353mm)
TRANSISTOR COUNT: 1569 SUBSTRATE CONNECTED TO GND
________________________________________________________Package Information
DIM A A1 B C D E e H L INCHES MAX MIN 0.104 0.094 0.011 0.004 0.017 0.011 0.012 0.009 0.610 0.604 0.298 0.292 0.032 BSC 0.416 0.398 0.035 0.020 8 0 MILLIMETERS MIN MAX 2.39 2.64 0.10 0.28 0.30 0.44 0.23 0.32 15.34 15.49 7.42 7.57 0.80 BSC 10.10 10.57 0.51 0.89 0 8
21-0032A
E
H
D A
0.127mm 0.004in.
e
B
A1
C
L
36-PIN PLASTIC SHRINK SMALL-OUTLINE PACKAGE
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
28 __________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 (408) 737-7600 (c) 1994 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

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